Acoustic filter employing inductive coupling

ABSTRACT

Disclosed in one embodiment is filter circuitry having first and second paths extending between first and second nodes. The first path has a first inductor and a second inductor coupled in series between the first node and the second node, wherein the first inductor and the second inductor are positively coupled with one another, and a first common node is provided between the first inductor and the second inductor. First shunt acoustic resonators are coupled between the first common node and a fixed voltage node. The second path includes a third inductor and a fourth inductor coupled in series between the first node and the second node. The third inductor and the fourth inductor are negatively coupled with one another, and a second common node is provided between the third inductor and the fourth inductor. Second acoustic resonators are coupled between the second common node and a fixed voltage node.

RELATED APPLICATIONS

This application claims the benefit of provisional patent applicationSer. No. 62/394,824, filed Sep. 15, 2016, the disclosure of which ishereby incorporated herein by reference in its entirety.

FIELD OF THE DISCLOSURE

The present disclosure relates to acoustic filters that employ inductivecoupling.

BACKGROUND

Acoustic resonators, such as Surface Acoustic Wave (SAW) resonators andBulk Acoustic Wave (BAW) resonators, are used in many high-frequencycommunication applications. In particular, SAW resonators are oftenemployed in filter networks that operate at frequencies up to 1.8 GHz,and BAW resonators are often employed in filter networks that operate atfrequencies above 1.5 GHz. Such filters need to have flat passbands,have steep filter skirts and squared shoulders at the upper and lowerends of the passband, and provide excellent rejection outside of thepassband. SAW- and BAW-based filters also have relatively low insertionloss, tend to decrease in size as the frequency of operation increases,and are relatively stable over wide temperature ranges.

As such, SAW- and BAW-based filters are the filter of choice for many3rd Generation (3G) and 4th Generation (4G) wireless devices and aredestined to dominate filter applications for 5th Generation (5G)wireless devices. Most of these wireless devices support cellular,Wireless Fidelity (WiFi), Bluetooth, and/or near field communications onthe same wireless device and, as such, pose extremely challengingfiltering demands. While these demands keep raising the complexity ofwireless devices, there is a constant need to improve the performance ofacoustic resonators and filters that are based thereon.

To better understand acoustic resonators and the various terminologyassociated therewith, the following provides an overview of a BAWresonator. However, the concepts described herein may employ any type ofacoustic resonator, such as SAW-based resonators, and are not limited toBAW-based resonators. An exemplary BAW resonator 10 is illustrated inFIG. 1. The BAW resonator 10 generally includes a substrate 12, areflector 14 mounted over the substrate 12, and a transducer 16 mountedover the reflector 14. The transducer 16 rests on the reflector 14 andincludes a piezoelectric layer 18, which is sandwiched between a topelectrode 20 and a bottom electrode 22. The top and bottom electrodes 20and 22 may be formed of Tungsten (W), Molybdenum (Mo), Platinum (Pt), orlike material, and the piezoelectric layer 18 may be formed of AluminumNitride (AlN), Zinc Oxide (ZnO), or other appropriate piezoelectricmaterial. Although shown in FIG. 1 as each including a single layer, thepiezoelectric layer 18, the top electrode 20, and/or the bottomelectrode 22 may include multiple layers of the same material, multiplelayers in which at least two layers are different materials, or multiplelayers in which each layer is a different material.

The BAW resonator 10 is divided into an active region 24 and an outsideregion 26. The active region 24 generally corresponds to the section ofthe BAW resonator 10 where the top and bottom electrodes 20 and 22overlap and also includes the layers below the overlapping top andbottom electrodes 20 and 22. The outside region 26 corresponds to thesection of the BAW resonator 10 that surrounds the active region 24.

For the BAW resonator 10, applying electrical signals across the topelectrode 20 and the bottom electrode 22 excites acoustic waves in thepiezoelectric layer 18. These acoustic waves primarily propagatevertically. A primary goal in BAW resonator design is to confine thesevertically propagating acoustic waves in the transducer 16. Acousticwaves traveling upward are reflected back into the transducer 16 by theair-metal boundary at the top surface of the top electrode 20. Acousticwaves traveling downward are reflected back into the transducer 16 bythe reflector 14 or by an air cavity, which is provided just below thetransducer in a Film BAW Resonator.

The reflector 14 is typically formed by a stack of reflector layers (RL)28, which alternate in material composition to produce a significantreflection coefficient at the junction of adjacent reflector layers 28.Typically, the reflector layers 28 alternate between materials havinghigh and low acoustic impedances, such as Tungsten (W) and SiliconDioxide (SiO₂). While only five reflector layers 28 are illustrated inFIG. 1, the number of reflector layers 28 and the structure of thereflector 14 varies from one design to another.

The magnitude (Z) and phase (φ) of the electrical impedance as afunction of the frequency for a relatively ideal BAW resonator 10 isprovided in FIG. 2. The magnitude (Z) of the electrical impedance isillustrated by the solid line, whereas the phase (φ) of the electricalimpedance is illustrated by the dashed line. A unique feature of the BAWresonator 10 is that it has both a resonance frequency and ananti-resonance frequency. The resonance frequency is typically referredto as the series resonance frequency (f_(s)), and the anti-resonancefrequency is typically referred to as the parallel resonance frequency(f_(p)). The series resonance frequency (f_(s)) occurs when themagnitude of the impedance, or reactance, of the BAW resonator 10approaches zero. The parallel resonance frequency (f_(p)) occurs whenthe magnitude of the impedance, or reactance, of the BAW resonator 10peaks at a significantly high level. In general, the series resonancefrequency (f_(s)) is a function of the thickness of the piezoelectriclayer 18 and the mass of the bottom and top electrodes 20 and 22.

For the phase, the BAW resonator 10 acts like an inductance thatprovides a 90° phase shift between the series resonance frequency(f_(s)) and the parallel resonance frequency (f_(p)). In contrast, theBAW resonator 10 acts like a capacitance that provides a −90° phaseshift below the series resonance frequency (f_(s)) and above theparallel resonance frequency (f_(p)). The BAW resonator 10 presents avery low, near zero, resistance at the series resonance frequency(f_(s)) and a very high resistance at the parallel resonance frequency(f_(p)). The electrical nature of the BAW resonator 10 lends itself tothe realization of a very high Q (quality factor) inductance over arelatively short range of frequencies, which has proved to be verybeneficial in high-frequency filter networks, especially those operatingat frequencies around 1.8 GHz and above.

Unfortunately, the phase (φ) curve of FIG. 2 is representative of anideal phase curve. In reality, approaching this ideal is challenging. Atypical phase curve for the BAW resonator 10 of FIG. 1 is illustrated inFIG. 3A. Instead of being a smooth curve, the phase curve of FIG. 3Aincludes ripple below the series resonance frequency (f_(s)), betweenthe series resonance frequency (f_(s)) and the parallel resonancefrequency (f_(p)), and above the parallel resonance frequency (f_(p)).The ripple is the result of spurious modes, which are caused by spuriousresonances that occur in corresponding frequencies. While the vastmajority of the acoustic waves in the BAW resonator 10 propagatevertically, various boundary conditions about the transducer 16 resultin the propagation of lateral (horizontal) acoustic waves, which arereferred to as lateral standing waves. The presence of these lateralstanding waves reduces the potential Q associated with the BAW resonator10.

As illustrated in FIG. 4, a border (BO) ring 30 is formed on or withinthe top electrode 20 to suppress certain of the spurious modes. Thespurious modes that are suppressed by the BO ring 30 are those above theseries resonance frequency (f_(s)), as highlighted by circles A and B inthe phase curve of FIG. 3B. Circle A shows a suppression of the ripple,and thus of the spurious mode, in the passband of the phase curve, whichresides between the series resonance frequency (f_(s)) and the parallelresonance frequency (f_(p)). Circle B shows suppression of the ripple,and thus of the spurious modes, above the parallel resonance frequency(f_(p)). Notably, the spurious mode in the upper shoulder of thepassband, which is just below the parallel resonance frequency f_(p),and the spurious modes above the passband are suppressed, as evidencedby the smooth or substantially ripple free phase curve between theseries resonance frequency (f_(s)) and the parallel resonance frequency(f_(p)) and above the parallel resonance frequency (f_(p)).

The BO ring 30 corresponds to a mass loading of the portion of the topelectrode 20 that extends about the periphery of the active region 24.The BO ring 30 may correspond to a thickened portion of the topelectrode 20 or the application of additional layers of an appropriatematerial over the top electrode 20. The portion of the BAW resonator 10that includes and resides below the BO ring 30 is referred to as a BOregion 32. Accordingly, the BO region 32 corresponds to an outer,perimeter portion of the active region 24 and resides inside of theactive region 24.

While the BO ring 30 is effective at suppressing spurious modes abovethe series resonance frequency (f_(s)), the BO ring 30 has little or noimpact on those spurious modes below the series resonance frequency(f_(s)), as shown by the ripples in the phase curve below the seriesresonance frequency (f_(s)) in FIG. 3B. A technique referred to asapodization is often used to suppress the spurious modes that fall belowthe series resonance frequency (f_(s)).

Apodization tries to avoid, or at least significantly reduce, anylateral symmetry in the BAW resonator 10, or at least in the transducer16 thereof. The lateral symmetry corresponds to the footprint of thetransducer 16, and avoiding the lateral symmetry corresponds to avoidingsymmetry associated with the sides of the footprint. For example, onemay choose a footprint that corresponds to a pentagon instead of asquare or rectangle. Avoiding symmetry helps reduce the presence oflateral standing waves in the transducer 16. Circle C of FIG. 3Cillustrates the effect of apodization in which the spurious modes belowthe series resonance frequency (f_(s)) are suppressed, as evidence bythe smooth or substantially ripple free phase curve below the seriesresonance frequency (f_(s)). Assuming no BO ring 30 is provided, one canreadily see in FIG. 3C that apodization fails to suppress those spuriousmodes above the series resonance frequency (f_(s)). As such, the typicalBAW resonator 10 employs both apodization and the BO ring 30.

As noted previously, BAW resonators 10 are often used in filter networksthat operate at high frequencies and require high Q values. A basicladder network 40 is illustrated in FIG. 5A. The ladder network 40includes two series resonators B_(SER) and two shunt resonators B_(SH),which are arranged in a traditional ladder configuration. Typically, theseries resonators B_(SER) have the same or similar first frequencyresponse, and the shunt resonators B_(SH) have the same or similarsecond frequency response, which is different from the first frequencyresponse, as shown in FIG. 5B. In many applications, the shuntresonators B_(SH) are detuned versions of the series resonators B_(SER).As a result, the frequency responses for the series resonators B_(SER)and the shunt resonators B_(SH) are generally very similar, yet shiftedrelative to one another such that the parallel resonance frequency(f_(p,SH)) of the shunt resonators approximates the series resonancefrequency (f_(s,SER)) of the series resonators B_(SER). Note that theseries resonance frequency (f_(s,SH)) of the shunt resonators B_(SH) isless than the series resonance frequency (f_(s,SER)) of the seriesresonators B_(SER). The parallel resonance frequency (f_(p,SH)) of theshunt resonators B_(SH) is less than the parallel resonance frequency(f_(p,SER)) of the series resonators B_(SER).

FIG. 5C is associated with FIG. 5B and illustrates the response of theladder network 40. The series resonance frequency (f_(s,SH)) of theshunt resonators B_(SH) corresponds to the low side of the passband'sskirt (phase 2), and the parallel resonance frequency (f_(p,SER)) of theseries resonators B_(SER) corresponds to the high side of the passband'sskirt (phase 4). The substantially aligned series resonance frequency(f_(s,SER)) of the series resonators B_(SER) and the parallel resonancefrequency (f_(p,SH)) of the shunt resonators B_(SH) fall within thepassband. FIGS. 6A through 6E provide circuit equivalents for the fivephases of the response of the ladder network 40. During the first phase(phase 1, FIGS. 5C, 6A), the ladder network 40 functions to attenuatethe input signal. As the series resonance frequency (f_(s,SH)) of theshunt resonators B_(SH) is approached, the impedance of the shuntresonators B_(SH) drops precipitously such that the shunt resonatorsB_(SH) essentially provide a short to ground at the series resonancefrequency (f_(s,SH)) of the shunt resonators (phase 2, FIGS. 5C, 6B). Atthe series resonance frequency (f_(s,SH)) of the shunt resonators B_(SH)(phase 2), the input signal is essentially blocked from the output ofthe ladder network 40.

Between the series resonance frequency (f_(s,SH)) of the shuntresonators B_(SH) and the parallel resonance frequency (f_(p,SER)) ofthe series resonators B_(SER), which corresponds to the passband, theinput signal is passed to the output with relatively little or noattenuation (phase 3, FIGS. 5C, 6C). Within the passband, the seriesresonators B_(SER) present relatively low impedance, whereas the shuntresonators B_(SH) present relatively high impedance, wherein thecombination of the two leads to a flat passband with steep low- andhigh-side skirts. As the parallel resonance frequency (f_(p,SER)) of theseries resonators B_(SER) is approached, the impedance of the seriesresonators B_(SER) becomes very high, such that the series resonatorsB_(SER) essentially present themselves as open at the parallel resonancefrequency (f_(p,SER)) of the series resonators (phase 4, FIGS. 5C, 6D).At the parallel resonance frequency (f_(p,SER)) of the series resonatorsB_(SER) (phase 4), the input signal is again essentially blocked fromthe output of the ladder network 40.

During the final phase (phase 5, FIGS. 5C, 6E), the ladder network 40functions to attenuate the input signal, in a similar fashion to thatprovided in phase 1. As the parallel resonance frequency (f_(p,SER)) ofthe series resonators B_(SER) is passed, the impedance of the seriesresonators B_(SER) decreases and the impedance of the shunt resonatorsB_(SH) normalizes. Thus, the ladder network 40 functions to provide ahigh Q passband between the series resonance frequency (f_(s,SH)) of theshunt resonators B_(SH) and the parallel resonance frequency (f_(p,SER))of the series resonators B_(SER). The ladder network 40 providesextremely high attenuation at both the series resonance frequency(f_(s,SH)) of the shunt resonators B_(SH) and the parallel resonancefrequency (f_(p,SER)) of the series resonators. The ladder network 40provides good attenuation below the series resonance frequency(f_(s,SH)) of the shunt resonators B_(SH) and above the parallelresonance frequency (f_(p,SER)) of the series resonators B_(SER). Asnoted previously, there is a constant need to improve the performance ofacoustic resonators and filters that are based thereon.

SUMMARY

Disclosed in one embodiment is filter circuitry having first and secondpaths extending between first and second nodes. The first path has afirst inductor and a second inductor coupled in series between the firstnode and the second node, wherein the first inductor and the secondinductor are positively coupled with one another, and a first commonnode is provided between the first inductor and the second inductor.First shunt acoustic resonators are coupled between the first commonnode and a fixed voltage node. The second path includes a third inductorand a fourth inductor coupled in series between the first node and thesecond node. The third inductor and the fourth inductor are negativelycoupled with one another, and a second common node is provided betweenthe third inductor and the fourth inductor. Second acoustic resonatorsare coupled between the second common node and a fixed voltage node. Thefilter response of the filter circuitry is one of a passband, astopband, or a multiple passband/stopband provided between the firstnode and the second node.

Disclosed in another embodiment is filter circuitry having an inputnode, a first node, a second node, an output node, and a fixed voltagenode. First acoustic resonators are coupled in parallel with one anotherbetween the input node and the first node. A first inductor is coupledbetween the first node and the fixed voltage node, such that the firstacoustic resonators are coupled in series with the first inductorbetween the input node and the fixed voltage node. Second acousticresonators are coupled in parallel with one another between the secondnode and the output node. A second inductor is coupled between thesecond node and the fixed voltage node, such that the second acousticresonators are coupled in series with the second inductor between theoutput node and the fixed voltage node, wherein the first inductor andthe second inductor are negatively coupled with one another.

The filter circuit may also have a third node and a fourth node, whereinthird acoustic resonators are coupled in parallel with one anotherbetween the input node and the first node. A third inductor is coupledbetween the third node and the fixed voltage node, such that the thirdplurality of acoustic resonators is coupled in series with the thirdinductor between the input node and the fixed voltage node. Fourthacoustic resonators are coupled in parallel with one another between thefourth node and the output node. A fourth inductor is coupled betweenthe fourth node and the fixed voltage node, such that the fourthacoustic resonators are coupled in series with the fourth inductorbetween the output node and the fixed voltage node, wherein the thirdinductor and the fourth inductor are positively coupled with oneanother.

A differential power amplifier circuit is provided in yet anotherembodiment. The differential power amplifier circuit includes a firstpower amplifier having a first input node and a first output node and asecond power amplifier having a second input node and a second outputnode. A first transformer includes a primary side coupled between thefirst output node and the second output node and a secondary side formedfrom a first inductor and a second inductor coupled in series between afirst node and a second node, wherein a common node is provided betweenthe first inductor and the second inductor. At least one series acousticresonator is coupled between the first node and the second node, whereinat least one main series resonance is provided between the first nodeand the second node at a main resonance frequency through the at leastone series acoustic resonator. A first shunt acoustic resonator iscoupled between the common node and a fixed voltage node. A second shuntacoustic resonator is coupled between the common node and the fixedvoltage node, wherein a differential input is provided across the firstinput node and the second input node, and a single-ended output signalis provided at the second node.

Those skilled in the art will appreciate the scope of the presentdisclosure and realize additional aspects thereof after reading thefollowing detailed description of the preferred embodiments inassociation with the accompanying drawing figures.

BRIEF DESCRIPTION OF THE DRAWING FIGURES

The accompanying drawing figures incorporated in and forming a part ofthis specification illustrate several aspects of the disclosure and,together with the description, serve to explain the principles of thedisclosure.

FIG. 1 illustrates a conventional Bulk Acoustic Wave (BAW) resonator.

FIG. 2 is a graph of the magnitude and phase of impedance over frequencyresponses as a function of frequency for an ideal BAW resonator.

FIGS. 3A-3C are graphs of phase responses for various BAW resonatorconfigurations.

FIG. 4 illustrates a conventional BAW resonator with a border ring.

FIG. 5A is a schematic of a conventional ladder network.

FIGS. 5B and 5C are graphs of a frequency response for BAW resonators inthe conventional ladder network of FIG. 5A and a frequency response forthe conventional ladder network of FIG. 5A.

FIGS. 6A-6E are circuit equivalents for the ladder network of FIG. 5A atthe frequency points 1, 2, 3, 4, and 5, which are identified in FIG. 5C.

FIG. 7 is a block diagram of a mobile terminal according to oneembodiment.

FIG. 8 is a schematic of an RF front-end according to a firstembodiment.

FIG. 9 illustrates a rigid PCB and a flexible PCB coupled together bymultiple coaxial cables according to a first embodiment.

FIG. 10 is a schematic of an RF front-end according to a secondembodiment.

FIG. 11 illustrates an acoustic resonator in parallel with acompensation circuit, which includes a single shunt acoustic resonator.

FIG. 12 is a graph that illustrates exemplary frequency responses forthe acoustic resonator, compensation circuit, and overall circuit ofFIG. 11.

FIG. 13 illustrates an acoustic resonator in parallel with acompensation circuit, which includes at least two shunt acousticresonators, according to a first embodiment.

FIG. 14 is a graph that illustrates exemplary frequency responses forthe acoustic resonator, compensation circuit, and overall circuit ofFIG. 13.

FIG. 15 is a graph that compares actual frequency responses of theoverall circuits of FIGS. 11 and 13.

FIG. 16 illustrates a plurality of parallel acoustic resonators inparallel with a compensation circuit, which includes at least two shuntacoustic resonators, according to a second embodiment.

FIG. 17 is a graph that illustrates first exemplary frequency responsesfor the acoustic resonator, compensation circuit, and overall circuit ofFIG. 16.

FIG. 18 is a graph that illustrates second exemplary frequency responsesfor the acoustic resonator, compensation circuit, and overall circuit ofFIG. 16.

FIGS. 19A through 19D illustrate transformation of the T-circuitimpedance architecture of the compensation circuit of FIG. 13 to a π(pi) impedance model.

FIG. 20 illustrates the overall circuit of FIG. 13 using the π (pi)impedance model of FIG. 19D.

FIG. 21 is a graph illustrating the overall shunt impedance, Zres,according to one embodiment.

FIG. 22 is a graph illustrating the series equivalent impedance, ZA,according to one embodiment.

FIGS. 23A and 23B are graphs over different frequency rangesillustrating the absolute or magnitude of series impedance ZS, theseries equivalent impedance ZA, and overall series impedance ZAs,according to one embodiment.

FIG. 24 is an alternative filter configuration according to oneembodiment.

FIG. 25 is an alternative filter configuration according to oneembodiment.

FIG. 26 is an alternative filter configuration according to oneembodiment.

FIG. 27 is an alternative filter configuration according to oneembodiment.

FIG. 28 is an alternative filter configuration according to oneembodiment.

FIG. 29 is differential amplifier circuit that employs the conceptsprovided herein.

FIG. 30 illustrates front end circuitry for a communication device thatemploys the concepts provided herein according to a first embodiment.

FIG. 31 illustrates front end circuitry for a communication device thatemploys the concepts provided herein according to a first embodiment.

DETAILED DESCRIPTION

The embodiments set forth below represent the necessary information toenable those skilled in the art to practice the embodiments andillustrate the best mode of practicing the embodiments. Upon reading thefollowing description in light of the accompanying drawing figures,those skilled in the art will understand the concepts of the disclosureand will recognize applications of these concepts not particularlyaddressed herein. It should be understood that these concepts andapplications fall within the scope of the disclosure and theaccompanying claims.

It will be understood that, although the terms first, second, etc. maybe used herein to describe various elements, these elements should notbe limited by these terms. These terms are only used to distinguish oneelement from another. For example, a first element could be termed asecond element, and similarly, a second element could be termed a firstelement, without departing from the scope of the present disclosure. Asused herein, the term “and/or” includes any and all combinations of oneor more of the associated listed items.

It will be understood that when an element such as a layer, region, orsubstrate is referred to as being “on” or extending “onto” anotherelement, it can be directly on or extend directly onto the other elementor intervening elements may also be present. In contrast, when anelement is referred to as being “directly on” or extending “directlyonto” another element, there are no intervening elements present.Likewise, it will be understood that when an element such as a layer,region, or substrate is referred to as being “over” or extending “over”another element, it can be directly over or extend directly over theother element or intervening elements may also be present. In contrast,when an element is referred to as being “directly over” or extending“directly over” another element, there are no intervening elementspresent. It will also be understood that when an element is referred toas being “connected” or “coupled” to another element, it can be directlyconnected or coupled to the other element or intervening elements may bepresent. In contrast, when an element is referred to as being “directlyconnected” or “directly coupled” to another element, there are nointervening elements present.

Relative terms such as “below” or “above” or “upper” or “lower” or“horizontal” or “vertical” may be used herein to describe a relationshipof one element, layer, or region to another element, layer, or region asillustrated in the figures. It will be understood that these terms andthose discussed previously are intended to encompass differentorientations of the device in addition to the orientation depicted inthe figures.

The terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting of the disclosure.As used herein, the singular forms “a,” “an,” and “the” are intended toinclude the plural forms as well, unless the context clearly indicatesotherwise. It will be further understood that the terms “comprises,”“comprising,” “includes,” and/or “including” when used herein specifythe presence of stated features, integers, steps, operations, elements,and/or components but do not preclude the presence or addition of one ormore other features, integers, steps, operations, elements, components,and/or groups thereof.

Unless otherwise defined, all terms (including technical and scientificterms) used herein have the same meaning as commonly understood by oneof ordinary skill in the art to which this disclosure belongs. It willbe further understood that terms used herein should be interpreted ashaving a meaning that is consistent with their meaning in the context ofthis specification and the relevant art and will not be interpreted inan idealized or overly formal sense unless expressly so defined herein.

As defined herein, the term “coupled” without being preceded with theadjective “acoustically” refers to an electrical coupling as opposed toan acoustic coupling. The term “acoustically coupled” refers to anacoustic coupling as opposed to an electrical coupling. Further, thephrase “about the same as” when referring to the series resonancefrequency of two or more devices means that the series resonancefrequencies of the devices are within 0.1% of each other.

While the concepts provided herein are applicable to varioustechnologies, these concepts are particularly useful in mobileterminals, such as mobile telephones, tablets, computers, and like smartdevices. The following provides an overview of such devices. Today'smobile terminals must communicate using different communicationtechnologies in different bands, which vary significantly in bothbandwidth and frequency. To further complicate matters, data rates areever increasing and the there is a need to transmit and receive overthese different bands at the same time. As a result, mobile terminalshave very complicated front-end configurations and are starting toemploy multiple input, multiple output (MIMO) transmission and receptiontechnology, which requires the use of multiple antennas.

FIG. 7 is a block diagram of a mobile terminal 42 that incorporates fourantennas: a primary antenna A1, a secondary antenna A2, a tertiaryantenna A3, and a quaternary antenna A4. The mobile terminal 42generally includes control circuitry 44, which is associated with a userinterface (I/F) 46, and radio frequency (RF) circuitry 48. The userinterface 46 may include microphones, speakers, keypads, touchscreens,displays, and the like. The RF circuitry 48 may include baseband,transceiver, power amplifier, and switching circuitry, as will beappreciated by those skilled in the art.

In general, signals to be transmitted are provided by the RF circuitry48 to one or more of the antennas A1 through A4, and signals received byone or more of the antennas A1 through A4 are routed to the RF circuitry48 for demodulation and associated processing. The RF circuitry 48 maybe configured to facilitate any number of communications, includingfirst-, second-, third-, fourth-, and fifth-generation cellularcommunications; wireless local area network communications; Bluetoothcommunications; industrial, scientific, and medical communications; nearfield communications; and the like. Any of these communications may useMIMO for transmission, reception, or both, depending on the capabilitiesof the mobile terminal 42 and the systems with which the mobile terminal42 communicates.

Since mobile terminals 42 are relatively small, the multiple antennas A1through A4 used for MIMO are relatively close to one another. As aresult, the antennas A1 through A4 may interact with one another, and asa result, modify each other's radiation patterns, which generally altersthe antenna's radiation efficiency. With continued reference to FIG. 7,when the primary antenna A1 is used for transmission and the tertiaryantenna A3 is used for reception at the same time, the transmission fromantennae A1 may significantly degrade the ability to receive signals viaantenna A3, given the proximity of antenna A1 to antenna A3. Further,the secondary antenna A2 and the quaternary antenna A4 may also beaffected by transmissions from the primary antenna A1. As such, there isa need for a cost-effective and space-efficient technique to resolve, orat least significantly reduce, the impact that one antenna has onanother in devices such as the mobile terminal 42 illustrated in FIG. 7.

With reference to FIG. 8, a technique for addressing the foregoingissues is described. As illustrated, the RF circuitry 48 is associatedwith antennas A1 through A4. Each of the antennas is associated withantenna tuning circuitry 50, 54, 52, 56, respectively. In particular,antenna A1 is coupled to the RF circuitry 48 through coaxial cable 58.Antenna A3 is coupled to the RF circuitry 48 through coaxial cable 60and a MPMS filter 62; antenna A2 is coupled to the RF circuitry 48through coaxial cable 64 and MPMS filter 66; and antenna A4 is coupledto the RF circuitry 48 through coaxial cable 68 and MPMS filter 70.While MPMS filters 62, 66, and 70 are provided for antennas A3, A2, andA4, respectively, alternative embodiments may only employ MPMS filter62, given the proximity of antennas A1 and A3. In other embodiments, anappropriately configured MPMS filter (not illustrated) may also beprovided in association with antenna A1. In short, MPMS filters may beprovided for each antenna A1 through A4 or any combination thereof. Theantenna tuning circuitry 50, 52, 54, 56 are used for tuning impedancesassociated with the respective antennas A1, A3, A2, A4, as those skilledin the art will appreciate.

For the following description, MPMS filter 62 is described in detail;however, MPMS filter 66 and 70 may be similarly or identicallyconfigured, depending on the embodiment. Assume that the RF circuitry 48is configured to transmit RF signals in band X and band Y via antenna A1at the same or different times. Further assume that RF circuitry 48 isconfigured to receive RF signals and bands A, B, and C at the same ordifferent times. Given the proximity of antennas A1 and A3, transmittingin bands X or Y via antenna A1 would significantly affect the ability ofantenna A3 to receive RF signals in bands A, B, or C, in the absence ofMPMS filter 62. However, adding MPMS filter 62 in proximity to antennaA3 significantly reduces the impact that antenna A1 has on antenna A3.

MPMS filter 62 is a specially configured filter that has multiplepassbands and multiple stopbands, which are interleaved with oneanother, as illustrated in FIG. 8. In this example, passbands areprovided for bands A, B, and C and stopbands are provided for at leastbands X and Y. A stopband for band X is between passbands for bands Aand B, and a stopband for band Y is between passbands for bands B and C.In other words, stopbands are provided for the problematic bands thatare transmitted via antenna A1, and passbands are provided for the bandsto be received via antenna A3. The RF circuitry 48 may also transmitsignals in bands A, B, or C via antenna A3. Providing stopbands for anadjacent antenna's transmission bands and passbands for the selectedantenna's receive (and transmission) bands can significantly improve theperformance of both antennas. When multiple ones of the MPMS filters 62,66, 70 are employed, the passbands and stopbands may be the same ordifferent among the different MPMS filters 62, 66, 70, based on theproximity of the antennas A1-A4 and the communication bands used forcommunications by the mobile terminal 42.

In certain embodiments, at least two of the stopbands and/or passbandsprovided by one or more of the MPMS filters 62, 66, 70 reside entirelyabove 2 GHz and have a bandwidth of at least 20 MHz. In otherembodiments, at least two of the stopbands and/or passbands residebetween 2 GHz and 12 GHz and have a bandwidth of at least 20 MHz, 40MHz, 50 MHz, or 100 MHz. In select embodiments, at least one of thestopbands or passbands residing between two other stopbands or passbandshas a bandwidth of at least 100 MHz, 150 MHz, or 200 MHz. All, or atleast certain of, the stopbands may provide attenuation of at least 10dB, 20 dB, or 30 dB in each of the foregoing embodiments, depending onthe configuration of the MPMS filters 62, 66, 70.

With reference to FIG. 9, the mobile terminal 42 may employ multipleprinted circuit boards (PCBs) to implement the necessary electronics foroperation. Further, the various antennas A1-A4 may be spread about themobile terminal 42. These antennas A1-A4 may be implemented on or in ahousing H (illustrated in FIG. 7) of the mobile terminal 42, on thevarious PCBs, or a combination thereof. FIG. 9 illustrates a rigid PCB(R-PCB) and a flexible PCB (F-PCB), which are used to implement at leastpart of the electronics for the mobile terminal 42. In one embodiment,the rigid PCB R-PCB may be a traditional glass-reinforced multilayercircuit board, wherein the flexible PCB F-PCB is provided by a muchthinner, flexible substrate on which traces and components may be formedor mounted. The flexible PCB F-PCB has a flex factor of at least tentimes that of the rigid PCB R-PCB.

As illustrated, the control circuitry 44 and the RF circuitry 48 areimplemented in whole or in part on the rigid PCB R-PCB while the MPMSfilters 62, 66, 70 and the antenna tuning circuitry 50, 52, 54, 56 areimplemented on the flexible PCB F-PCB. The coaxial cables 58, 60, 64, 68connect the rigid PCB R-PCB and the flexible PCB F-PCB such that thetransmit/receive paths that extend between the RF circuitry 48 and therespective antennas A1, A2, A3, and A4 are provided by the combinationof the rigid PCB R-PCB, the flexible PCB F-PCB, and the coaxial cables58, 60, 64, 68. These transmit/receive paths extend to correspondingantenna ports AP1, AP2, AP3, AP4 of the flexible PCB F-PCB. The antennasA1, A2, A3, and A4 are connected to the antenna ports AP1, AP2, AP3,AP4, respectively, through cables, traces, and/or the like.

With reference to FIG. 10, a low-noise amplifier (LNA) 72 may beprovided between the MPMS filter 62 and the coaxial cable 60 to amplifythe filtered receive signals prior to the coaxial cable 60. The LNA 72may be provided along with the MPMS filter 62 on the flexible PCB F-PCB,wherein the RF circuitry 48 is provided on the rigid PCB R-PCB, and thecoaxial cable 60 connects the flexible PCB F-PCB and the rigid PCBR-PCB. The antenna tuning circuitry 50, 52 may also be provided on theflexible PCB F-PCB.

The following provides various filters that employ acoustic resonatorsand are capable of providing a filter response that includes multiplepassbands and stopbands. Some basics regarding the theory of operationare provided prior to describing the specific configurations, whichprovide the desired filter responses.

Turning now to FIG. 11, a series resonator B1 is shown coupled betweenan input node VP and an output node O/P. The series resonator B1 has aseries resonance frequency F_(s) and inherent capacitance, whichgenerally limits the bandwidth of filters that employ the seriesresonator B1. In the case of a Bulk Acoustic Wave (BAW) resonator, thecapacitance of the series resonator B1 is primarily caused by itsinherent structure, which looks and acts like a capacitor in partbecause the series resonator includes the top and bottom electrodes 20,22 (FIG. 1) that are separated by a dielectric piezoelectric layer 18.While BAW resonators are the focus of this and the following examples,other types of acoustic resonators, such as Surface Acoustic Wave (SAW)resonators, are equally applicable.

A compensation circuit 74 is coupled in parallel with the seriesresonator B1 and functions to compensate for some of the capacitancepresented by the series resonator B1. The compensation circuit 74includes two negatively coupled inductors L1, L2 and a shunt resonatorB2. The negatively coupled inductors L1, L2 are coupled in seriesbetween the input node I/P and the output node O/P, wherein a commonnode CN is provided between the negatively coupled inductors L1, L2. Thenegatively coupled inductors L1, L2 are magnetically coupled by acoupling factor K, wherein the dots illustrated in association with thenegatively coupled inductors L1, L2 indicate that the magnetic couplingis negative. As such, the negatively coupled inductors L1, L2 areconnected in electrical series and negatively coupled from a magneticcoupling perspective. As defined herein, two (or more) series-connectedinductors that are negatively coupled from a magnetic perspective areinductors that are:

-   -   connected in electrical series; and    -   the mutual inductance between the two inductors functions to        decrease the total inductance of the two (or more) inductors.

The shunt resonator B2 is coupled between the common node CN and ground,or other fixed voltage node.

To compensate for at least some of the capacitance of the seriesresonator B1, the compensation circuit 74 presents itself as a negativecapacitance within certain frequency ranges when coupled in parallelwith the series resonator B1. Since capacitances in parallel areadditive, providing a negative capacitance in parallel with the(positive) capacitance of the series resonator B1 effectively reducesthe capacitance of the series resonator B1. With the compensationcircuit 74, the series resonator B1 can actually function as a filter(instead of just a resonator) and provide a passband, albeit a fairlynarrow passband, instead of a more traditional resonator response (solidline of FIG. 2).

FIG. 12 graphically illustrates the frequency responses of the seriesresonator B1 (inside the block referenced B1), the compensation circuit74 (inside the block referenced 74), and the overall circuit in whichthe compensation circuit 74 is placed in parallel with the seriesresonator B1. As illustrated, the overall circuit provides a relativelynarrow passband. Further detail on this particular circuit topology canbe found in the co-assigned U.S. patent application Ser. No. 15/004,084,filed Jan. 22, 2016, and titled RF LADDER FILTER WITH SIMPLIFIEDACOUSTIC RF RESONATOR PARALLEL CAPACITANCE COMPENSATION, and U.S. patentapplication Ser. No. 14/757,651, filed Dec. 23, 2015, and titledSIMPLIFIED ACOUSTIC RF RESONATOR PARALLEL CAPACITANCE COMPENSATION, thedisclosures of which are incorporated herein by reference in theirentireties.

While beneficial in many applications, the narrow passband of thecircuit topology of FIG. 11 has its limitations. With the challenges ofmodern day communication systems, wider passbands and the ability toprovide multiple passbands within a given system are needed.Fortunately, applicants have discovered that certain modifications tothis topology provide significant and truly unexpected increases inpassband bandwidths and, in certain instances, the ability to generatemultiple passbands of the same or varying bandwidths in an efficient andeffective manner.

With reference to FIG. 13, a modified circuit topology is illustratedwherein the circuit topology of FIG. 11 is modified to include anadditional shunt resonator B3, which is coupled between the common nodeCN and ground. As such, a new compensation circuit 76 is created thatincludes the negatively coupled inductors L1 and L2, which have acoupling coefficient K, and at least two shunt resonators B2, B3. Thecompensation circuit 76 is coupled in parallel with the series resonatorB1. When the series resonance frequencies F_(s) of the shunt resonatorsB2, B3 are different from one another, unexpectedly wide bandwidthspassbands are achievable while maintaining a very flat passbands, steepskirts, and excellent cancellation of signals outside of the passbands.

FIG. 14 graphically illustrates the frequency responses of the seriesresonator B1 (inside the block referenced B1), the compensation circuit76 (inside the block referenced 76), and the overall circuit in whichthe compensation circuit 76 is placed in parallel with the seriesresonator B1. As illustrated, the overall circuit with the compensationcircuit 76 provides a much wider passband (FIG. 10) than the overallcircuitry with the compensation circuit 74 (FIG. 12).

While FIGS. 12 and 14 are graphical representations, FIG. 15 is anactual comparison of the frequency response of the overall circuit usingthe different compensation circuits 74, 76, wherein the overall circuitusing the compensation circuit 76 provides a significantly wider andbetter formed passband (solid line) than the overall circuit using thecompensation circuit 74 (dashed line).

As illustrated in FIG. 16, the concepts described herein not onlycontemplate the use of multiple shunt resonators B2, B3, which arecoupled between the common node CN and ground, but also multiple seriesresonators, such as series resonators B1 and B4, which are coupled inparallel with one another between the input node VP and the output nodeO/P. The series resonance frequencies F_(s) of the series resonators B1,B4 are different from one another, and the series resonance frequenciesF_(s) of the shunt resonators B2, B3 are also different from one anotherand different from those of the series resonators. While only two seriesresonators B1, B4 and two shunt resonators B2, B3 are illustrated, anynumber of these resonators may be employed depending on the applicationand the desired characteristics of the overall frequency response of thecircuit in which these resonators and associated compensation circuits76 are employed. While the theory of operation is described furthersubsequently, FIGS. 17 and 18 illustrate just two of the manypossibilities.

For FIG. 17, there are two series resonators B1, B4 and two shuntresonators B2, B3, with different and relatively dispersed seriesresonance frequencies F_(s). FIG. 17 graphically illustrates thefrequency responses of the combination of the two series resonators B1,B4 (inside the block referenced BX), the compensation circuit 76 withtwo shunt resonators B2, B3 (inside the block referenced 76), and theoverall circuit in which the compensation circuit 76 is placed inparallel with the series resonator B1. As illustrated, the overallcircuit in this configuration has the potential to provide a passbandthat is even wider than that for the embodiment of FIGS. 13 and 14. Forexample, passbands of greater than 100 MHz, 150 MHz, 175 MHz, and 200MHz are contemplated at frequencies at or above 1.5 GHz, 1.75 GHz, and 2GHz.

In other words, center-frequency-to-bandwidth ratios (fc/BW*100) of 3.5%to 9%, 12%, or greater are possible, wherein fc is the center frequencyof the passband and BW is the bandwidth of the passband. If multiplepassbands are provided, BW may encompass all of the provided passbands.Further, when multiple passbands are provided, the passbands may havethe same or different bandwidths or center-frequency-to-bandwidthratios. For example, one passband may have a relatively largecenter-frequency-to-bandwidth ratio, such as 12%, and a second passbandmay have a relatively small center-frequency-to-bandwidth ratio, such as2%. Alternatively, multiple ones of the passbands may have a bandwidthof 100 MHz, or multiple ones of the passbands may have generally thesame center-frequency-to-bandwidth ratios. In the latter case, thebandwidths of the passbands may inherently be different from oneanother, even though the center-frequency-to-bandwidth ratios are thesame.

For FIG. 18, there are four series resonators, which are coupled inparallel with one another (not shown), and two shunt resonators (notshown) with different and more widely dispersed series resonancefrequencies F_(s). FIG. 18 graphically illustrates the frequencyresponses of the combination of the four series resonators (inside theblock referenced BX), the compensation circuit 76 with two shuntresonators B2, B3 (inside the block referenced 76), and the overallcircuit in which the compensation circuit 76 is placed in parallel withthe series resonator B1. As illustrated, the overall circuit in thisconfiguration provides multiple passbands, which are separated by astopband. In this embodiment, two passbands are provided; however, thenumber of passbands may exceed two. The number of passbands in thebandwidth of each of the passbands is a function of the number of shuntand series resonators B1-B4 and the series resonance frequencies F_(s)thereof.

The theory of the compensation circuit 76 follows and is described inassociation with FIGS. 19A through 19D and FIG. 20. With reference toFIG. 19A, assume the compensation circuit 76 includes the two negativelycoupled inductors L1, L2, which have an inductance value L, and two ormore shunt resonators BY, which have an overall shunt impedance Zrespresented between the common node CN and ground. While the inductancevalues L of the negatively coupled inductors L1, L2 are described asbeing the same, these values may differ depending on the application.Also assume that the one or more series resonators BX present an overallseries impedance ZS.

As shown in FIG. 19B, the two negatively coupled and series-connectedinductors L1, L2 (without Zres) can be modeled as a T-network of threeinductors L3, L4, and L5, wherein series inductors L3 and L4 areconnected in series and have a value of L(1+K), and shunt inductor L5has a value of −L*K, where K is a coupling factor between the negativelycoupled inductors L1, L2. Notably, the coupling factor K is a positivenumber between 0 and 1. Based on this model, the overall impedance ofthe compensation circuit 76 is modeled as illustrated in FIG. 19C,wherein the shunt impedance Zres is coupled between the shunt inductorL5 and ground. The resulting T-network, as illustrated in FIG. 19C, canbe transformed into an equivalent π (pi) network, as illustrated in FIG.19D.

The π network of FIG. 19D can be broken into a series equivalentimpedance ZA and two shunt equivalent impedances ZB. The seriesequivalent impedance ZA is represented by two series inductances ofvalue L*(1+K), where K>0, and a special “inversion” impedance Zinv. Theinversion impedance Zinv is equal to [L(1+K)ω]²/[Zres−jLKω], where ω=2πfand f is the frequency. As such, the series equivalent impedance ZAequals j*2*L(1+K)ω+Zinv and is coupled between the input node I/O andthe output node O/P. Each of the two shunt equivalent impedances ZB isrepresented by an inductor of value L(1−K) in series with two overallshunt impedances Zres.

Notably, the series equivalent impedance ZA has a negative capacitorbehavior at certain frequencies at which broadband cancellation isdesired and has series resonance at multiple frequencies. In general,the series equivalent impedance ZA has a multiple bandpass-bandstopcharacteristic, in that the series equivalent impedance ZA passes somefrequencies and stops others. When the series equivalent impedance ZA isplaced in parallel with the series impedance ZS of the series resonatorsBX, which can also have a multiple bandpass-bandstop characteristic, abroadband filter or a filter with multiple passbands may be created.

FIG. 20 illustrates the series impedance ZS of the series resonators BXin parallel with the series equivalent impedance ZA of the compensationcircuit 76. The overall series impedance ZAs represents the seriesimpedance ZS in parallel with the series equivalent impedance ZA. Thetwo shunt equivalent impedances ZB are respectively coupled between theinput port I/P and ground and the output port O/P and ground. Theprimary focus for the following discussion relates to the seriesequivalent impedance ZA and its impact on the series impedance ZS whenthe series equivalent impedance ZA is placed in parallel with the seriesimpedance ZS.

As noted previously, the series equivalent impedance ZA provides twoprimary functions. The first provides a negative capacitive behavior,and the second provides one or more additional series resonances betweenthe input node I/P and the output node O/P. These additional seriesresonances are provided through the series equivalent impedance ZA andare in addition to any series resonances that are provided through theseries impedance ZS of the series resonators BX. To help explain thebenefits and concept of the negative capacitive behavior provided by theseries equivalent impedance ZA, normal capacitive behavior isillustrated in association with the overall shunt impedance Zres, whichis provided by the shunt resonators BY. FIG. 21 graphs the absolute(magnitude) and imaginary components of the overall shunt impedanceZres, which is formed by two shunt resonators BY coupled in parallelwith one another.

The series resonance frequency F_(s) for each of the two shuntresonators BY occurs when the absolute impedance (abs(Zres)) is at ornear zero. Since there are two shunt resonators BY, the absoluteimpedance (abs(Zres)) is at or near zero at two frequencies, and assuch, there are two series resonance frequencies F_(s). The parallelresonance frequencies F_(p) occur when the imaginary component(imag(Zres)) peaks. Again, since there are two shunt resonators BY,there are two series resonance frequencies F_(s) provided by the overallshunt impedance Zres.

Whenever the imaginary component (imag(Zres)) of the overall shuntimpedance Zres is less than zero, the overall shunt impedance Zres has acapacitive behavior. The capacitive behavior is characterized in thatthe reactance of the overall shunt impedance Zres is negative anddecreases as frequency increases, which is consistent with capacitivereactance, which is represented by 1/jωC. The graph of FIG. 21identifies three regions within the impedance response of the overallshunt impedance Zres that exhibit capacitive behavior.

Turning now to FIG. 22, the series equivalent impedance ZA isillustrated over the same frequency range as that of the overall shuntimpedance

Zres, illustrated in FIG. 21. The series equivalent impedance ZA has twoseries resonance frequencies F_(s), which occur when the absoluteimpedance (abs(ZA)) is at or near zero. The two series resonancefrequencies F_(s) for the series equivalent impedance ZA are differentfrom each other and slightly different from those for the overall shuntimpedance Zres. Further, the number of series resonance frequenciesF_(s) generally corresponds to the number of shunt resonators BY in thecompensation circuit 76, assuming the series resonance frequencies F_(s)are different from one another.

Interestingly, the imaginary component (imag(ZA)) of the seriesequivalent impedance ZA is somewhat inverted with respect to that of theoverall shunt impedance Zres. Further, the imaginary component(imag(ZA)) of the series equivalent impedance ZA has a predominantlypositive reactance. During the portions at which the imaginary component(imag(ZA)) is positive, the reactance of the series equivalent impedanceZA again decreases as frequency increases, which is indicative ofcapacitive behavior. However, the reactance is positive, whereastraditional capacitive behavior presents a negative reactance. Thisphenomenon is referred to as negative capacitive behavior. Thoseportions of the imaginary component (imag(ZA)) of the series equivalentimpedance ZA that are positive and thus exhibit negative capacitivebehavior are highlighted in the graph of FIG. 22.

The negative capacitive behavior of the series equivalent impedance ZAfor the compensation circuit 76 is important, because when the seriesequivalent impedance ZA is placed in parallel with the series impedanceZS, the effective capacitance of the overall circuit is reduced.Reducing the effective capacitance of the overall circuit shifts theparallel resonance frequency F_(p) of the series impedance ZS higher inthe frequency range, which is described subsequently, and significantlyincreases the available bandwidth for passbands while providingexcellent out-of-band rejection.

An example of the benefit is illustrated in FIGS. 23A and 23B. Thethicker solid line, which is labeled abs(VG), represents the frequencyresponse of the overall circuit illustrated in FIG. 16, wherein thereare two series resonators BX and two shunt resonators BY in thecompensation circuit 76. The frequency response has two well-definedpassbands, which are separated by a stopband. The frequency responseabs(VG) of the overall circuit generally corresponds to the inverse ofthe overall series impedance ZAs, which again represents the seriesimpedance Zs in parallel with the series equivalent impedance ZA, asprovided in FIG. 20.

Notably, the parallel resonance frequencies F_(p)(ZS) of the seriesimpedance ZS, in isolation, fall in the middle of the passbands offrequency response abs(VG) of the overall circuit. If the parallelresonance frequencies F_(p)(ZS) of the series impedance ZS remained atthese locations, the passbands would be severely affected. However, thenegative capacitive behavior of the series equivalent impedance ZAfunctions to shift these parallel resonance frequencies F_(p)(ZS) of theseries impedance ZS to a higher frequency and, in this instance, abovethe respective passbands. This is manifested in the resulting overallseries impedance ZAs, in which the only parallel resonance frequenciesF_(p)(ZAs) occur above and outside of the respective passbands. Anadditional benefit to having the parallel resonance frequenciesF_(p)(ZAs) occur outside of the respective passbands is the additionalcancellation of frequencies outside of the passbands. Plus, the overallseries impedance ZAs is lower than the series impedance ZS within therespective passbands.

A further contributor to the exemplary frequency response abs(VG) of theoverall circuit is the presence of the additional series resonancefrequencies F_(s), which are provided through the series equivalentimpedance ZA. These series resonance frequencies F_(s) are offset fromeach other and from those provided through the series impedance ZS. Theseries resonance frequencies F_(s) for the series equivalent impedanceZA in the series impedance ZS occur when the magnitudes of therespective impedances approach zero. The practical results are widerpassbands, steeper skirts for the passbands, and greater rejectionoutside of the passbands, as evidenced by the frequency response abs(VG)of the overall circuit.

Turning now to FIG. 24, an alternative filter circuit that employsinductive coupling is provided. In essence, two filter paths P1, P2extend between an input node I/P and an output node O/P. The firstfilter path P1 includes two inductors L11, L12, which are positivelycoupled with one another and connected in series between the input nodeI/P and the output node O/P. The coupling factor Ka between the inductorL11 and inductor L12 is between zero and one. The inductors L11, L12 areconnected at a common node CN10. A plurality of acoustic resonators B11through B13 are coupled in parallel with one another between the commonnode CN10 and ground (or other fixed voltage node). While only threeacoustic resonators B11 through B13 are shown, any number of acousticresonators may be employed.

The second filter path P2 is similarly configured to that of the firstfilter path P1, and includes two inductors L21, L22, which arenegatively coupled with one another and connected in series between theinput node I/P and the output node O/P. The coupling factor Ks betweenthe inductor L21 and inductor L22 is between zero and one and may be thesame or different from that of the coupling factor Ka. The inductorsL21, L22 are connected at a common node CN20. A plurality of acousticresonators B21 through B23 are coupled in parallel with one anotherbetween the common node CN20 and ground. While only three acousticresonators B21 through B23 are shown, any number of acoustic resonatorsmay be employed.

The response of the filter circuit may be provide bandpass, notch(stopband), or multiple bandpass/stopband responses, depending on theselection of series resonance frequencies f11, f12, f13 for the acousticresonators B11, B12, B13 of the first filter path P1 and the seriesresonance frequencies f21, f22, f23 of the acoustic resonators B21, B22,B23 of the second filter path P2. In one embodiment, the seriesresonance frequencies f11, f12, f13, f21, f22, f23 for the acousticresonators B11, B12, B13, B21, B22, B23 are different from one anotherand are located to provide the desired filter response.

Turning now to FIG. 25, another filter circuit F1 that employs inductivecoupling is provided. Again, the filter circuit F1 extends between aninput node I/P and an output node O/P. The filter circuit F1 has a firstpath P3 that extends between the input node I/P and ground (or otherfixed voltage node). The first path P3 includes two or more acousticresonators B31, B32, which are coupled in parallel with one another andin series with an inductor L31. The filter circuit F1 has a second pathP4 that extends between the output node O/P and ground (or other fixedvoltage node). The second path P4 includes two or more acousticresonators B41, B42, which are coupled in parallel with one another andin series with an inductor L32. The inductors L31, L32 are negativelycoupled and have a coupling factor of K. In one embodiment, the acousticresonators B31, B32 of the first path P3 have series resonancefrequencies that differ from one another, and the acoustic resonatorsB41, B42 have series resonance frequencies that differ from one another.However, each acoustic resonator B31, B32 of the first path P3 may havethe same series resonance frequency as that of one of the acousticresonators B41, B42 of the second path P4. For example, acousticresonators B31, B41 may have a first series resonance frequency, andacoustic resonators B32, B42 may have a second series resonancefrequency, which is different from the first series resonance frequency.The series resonance frequencies of the various acoustic resonators B31,B32, B41, and B42 may also be different from one another in alternativeembodiments. As with the embodiment of FIG. 24, the series resonancefrequencies chosen for the acoustic resonators B31, B32, B41, and B42 inthe first and second paths P3, P4 dictates the type of filter responsefor the filter circuit F1, which may include bandpass, notch (stopband),or multiple bandpass/stopband responses.

As illustrated in FIG. 26, a variant of the filter circuit F1 of FIG. 25may be placed in parallel with the filter circuit F1 in order to providea combined filter circuit. In essence, the filter circuit F1 is placedin parallel with filter circuit F2, wherein both filter circuits F1, F2extend between the input node I/P and the output node O/P.

Notably, filter circuit F2 employs positive inductive coupling asopposed to the negative inductive coupling of filter circuit F1. Thefilter circuit F2 has a first path P3′ that extends between the inputnode I/P and ground (or other fixed voltage node). The first path P3′includes two or more acoustic resonators B31′, B32′, which are coupledin parallel with one another and in series with an inductor L31′. Thefilter circuit F2 has a second path P4′ that extends between the outputnode O/P and ground (or other fixed voltage node). The second path P4′includes two or more acoustic resonators B41′, B42′, which are coupledin parallel with one another and in series with an inductor L32′. Theinductors L31′, L32′ are negatively coupled. In one embodiment, theacoustic resonators B31′, B32′ of the first path P3′ have seriesresonance frequencies that differ from one another, and the acousticresonators B41′, B42′ have series resonance frequencies that differ fromone another. However, each acoustic resonator B31′, B32′ of the firstpath P3′ may have the same series resonance frequency as that of one ofthe acoustic resonators B41′, B42′ of the second path P4′. For example,acoustic resonators B31′, B41′ may have a first series resonancefrequency, and acoustic resonators B32′, B42′ may have a second seriesresonance frequency, which is different from the first series resonancefrequency. The series resonance frequencies of the various acousticresonators B31′, B32′, B41′, and B42′ may also be different from oneanother in alternative embodiments. In most embodiments, the respectiveacoustic resonators B31, B32, B41, and B42 of the first filter circuitF1 have series resonance frequencies that differ from the seriesresonance frequencies of the acoustic resonators B31′, B32′, B41′, andB42′ of the second filter circuit F2. While only two filter circuits F1,F2 are illustrated, additional filter circuits may be provided inparallel with these filter circuits F1, F2. Regardless of configuration,the overall filter circuit may provide bandpass, notch (stopband), ormultiple bandpass/stopband responses, depending on the selection ofresonance frequencies.

Yet another embodiment is illustrated in FIG. 27. The illustrated filtercircuit is similar to that illustrated in FIG. 16. The difference isthat an inductive and/or capacitive element 80 is placed in series withone or more of the acoustic resonators B2, B3 of the compensationcircuit 76. Additionally, another inductive and/or capacitive element 82may be placed in series with one or more of the series acousticresonators B1, B4. The inductive and/or capacitive elements 80, 82provide additional variables for adjusting the filter response of thefilter circuit.

FIG. 28 provides an additional variant of the filter circuit of FIG. 16,wherein the acoustic resonators B2, B3 of the compensation circuit 76are replaced with an LC circuit. In particular, a compensation circuit76′ includes the two negatively coupled inductors L1, L2, as describedpreviously and an inductor L′ and a capacitor C′, which are coupled inparallel with one another between the common node CN and ground (orother fixed voltage note). If the coupling factor K is equal to orapproaches 1, extremely broadband passbands or stopbands, as well asmultiple passbands and stopbands spread over a broad frequency range,are possible.

In particular (with concurrent reference to FIG. 16 and the descriptionthereof), ZA=j*2*L*(1+K)*w−[L(1+K)*w]²/[Zres−j*L*K*w].

Zres can be expressed from the equation as function of ZA, such that:

Zres=[+j*L*K*w*ZA−L²w²*(1+K)*(1−K)]/[ZA−j*2*L*(1+K)*w], and

Zres=[+j*L*K*w*ZA−L²w²*(1−K²)]/[ZA−j*2*L*(1+K)*w].

If we take only the parallel C0 capacitor part of Zs (i.e., capacitanceof series resonators B1, B4), Zs_C0=−j/(C0*w).

To synthesize a parallel ZA impedance that cancels C0:

ZA=−Zs_C0=+j/(C0*w).

Thus, we can replace ZA in the foregoing Zres equation, as follows:

Zres to create negativeC0=[+j²*L*K*w/(C0*w)−L²w²*(1−K²)]/[+j/(C0*w−j*2*L*(1+K)*w];

Zres to create negativeC0=[−LK/C0−L²w²*(1−K²)]/(+j*[+1/(C0*w)−2*L*(1+K)*w]);

Zres to create negativeC0=+j[LK/C0+L²w²*(1−K²)]/[+1/(C0*w)−2*L*(1+K)*w].

As a result, Yres=1/Zres to create negativeC0=−j*[+1/(C0*w)−2*L*(1+K)*w]/[K/C0+L²w²*(1−K²)].

If we have K=1 (ideal coupling), Yres for K=1=−j/(L*w)+j*4*C0*w.

Thus, Yres is made of two parallel admittances, an inductor L′ of valueL and a capacitor C′ of value 4*C0 for the case of two negativelycoupled inductors of L value with K=1. The result is exceptionally broadbandwidths from a single filter structure that has very low insertionloss.

FIG. 29 illustrates a unique implementation of the filter circuitillustrated in FIG. 16 and a differential amplifier circuit. Thedifferential amplifier circuit includes differential power amplifiers84, 86, which drive a primary (winding) side of a center-tappedtransformer T1. The center tap of the primary side of transformer T1 iscoupled to a supply voltage Vcc and to ground through a capacitor C″.The secondary (winding) side of the transformer T1 provides thenegatively coupled inductors L1 and L2 for the compensation circuit 76.The acoustic resonators B2, B3 for the compensation circuit 76 arecoupled between the center tap of the secondary side of the transformerT1 and ground (or other fixed voltage note). The series acousticresonators B1, B4 of the compensation circuit 76 are effectively coupledacross the secondary side of the transformer T1. In operation, adifferential input signal (DIFF I/P) drives the differential poweramplifiers 84, 86 and a single-ended output signal (O/P) is presented toa downstream circuit element, such as another ladder network 88.

Another application for the concepts disclosed herein is illustrated inFIG. 30. A transmit signal (TX signal) is presented to a power amplifier90 and passed through a single, multiband filter 92 to an antenna nodeAN, which is coupled directly or indirectly to an antenna (not shown).The multiband filter 92 is configured according to one or more of theconcepts described previously, and in this example, is configured toprovide three distinct passbands, which are separated by stopbands andare centered about transmit band A, transmit band B, and transmit bandC, respectively. Multiple receive filters 94, 96, 98 are also coupled tothe antenna node AN and are used to separate the receive signals andreceive band A, receive band B, and receive band C. As such, the single,multiband filter 92 may be used in the transmit path, while multiplereceive filters 94, 96, 98 may be used in the receive path, and viceversa.

Another variant is illustrated in FIG. 31, wherein the transmit signal(TX signal) is presented to a first port of a first hybrid coupler 102.A second port of the first hybrid coupler 102 is coupled to ground (orother fixed voltage noted) through a resistor R_(C). The third andfourth ports of the first hybrid coupler 102 are respectively coupled tothe inputs of two multiband filters 92, which provide multiple distinctpassbands, separated by stop bands. As illustrated, the multibandfilters 92 provide passbands about transmit band A, transmit band B, andtransmit band C. The respective outputs of the multiband filters 92drive first and second ports of a second hybrid coupler 104. The thirdport of the second hybrid coupler 104 is coupled to multiple receivefilters 94, 96, 98, which are used to pass signals in receive band A,receive band B, and receive band C, respectively.

While the concepts disclosed herein are described in association with amobile terminal, these concepts are applicable to any type ofcommunication device that employs wireless communications. Those skilledin the art will recognize numerous modifications and other embodimentsthat incorporate the concepts described herein. These modifications andembodiments are considered to be within scope of the teachings providedherein and the claims that follow.

What is claimed is:
 1. Filter circuit comprising: a first node and asecond node; a first path coupled between the first node and the secondnode and comprising: a first inductor and a second inductor coupled inseries between the first node and the second node, wherein the firstinductor and the second inductor are positively coupled with one anotherand a first common node is provided between the first inductor and thesecond inductor; and a first plurality of shunt acoustic resonatorscoupled between the first common node and a fixed voltage node; a secondpath coupled between the first node and the second node and comprising:a third inductor and a fourth inductor coupled in series between thefirst node and the second node, wherein the third inductor and thefourth inductor are negatively coupled with one another and a secondcommon node is provided between the third inductor and the fourthinductor; and a second plurality of shunt acoustic resonators coupledbetween the second common node and a fixed voltage node, wherein afilter response of one of a passband, a stopband, a multiplepassband/stopband is provided between the first node and the secondnode.
 2. The filter circuit of claim 1 wherein each of the firstplurality of shunt acoustic resonators has a series resonance frequencythat is different from the series resonance frequency of the other ofthe first plurality of shunt acoustic resonators.
 3. The filter circuitof claim 2 wherein each of the second plurality of shunt acousticresonators has a series resonance frequency that is different from theseries resonance frequency of the other of the second plurality of shuntacoustic resonators.
 4. The filter circuit of claim 1 wherein: each ofthe second plurality of shunt acoustic resonators has a series resonancefrequency that is different from the series resonance frequency of theother of the second plurality of shunt acoustic resonators and from theseries resonance frequency of each of the first plurality of shuntacoustic resonators; and each of the first plurality of shunt acousticresonators has a series resonance frequency that is different from theseries resonance frequency of the other of the first plurality of shuntacoustic resonators and from the series resonance frequency of each ofthe second plurality of shunt acoustic resonators.
 5. The filter circuitof claim 1 wherein the filter response is the multiplepassband/stopband.
 6. The filter circuit of claim 1 wherein a firstcoupling coefficient between the first inductor and the second inductoris different from a second coupling coefficient between the thirdinductor and the fourth inductor.
 7. The filter circuit of claim 1wherein the first plurality of shunt acoustic resonators and the secondplurality of shunt acoustic resonators are bulk acoustic waveresonators.
 8. Filter circuit comprising: an input node, a first node, asecond node, an output node, and a fixed voltage node; a first pluralityof acoustic resonators that are coupled in parallel with one anotherbetween the input node and the first node; a first inductor coupledbetween the first node and the fixed voltage node such that the firstplurality of acoustic resonators is coupled in series with the firstinductor between the input node and the fixed voltage node; a secondplurality of acoustic resonators that are coupled in parallel with oneanother between the second node and the output node; and a secondinductor coupled between the second node and the fixed voltage node suchthat the second plurality of acoustic resonators is coupled in serieswith the second inductor between the output node and the fixed voltagenode, wherein the first inductor and the second inductor are negativelycoupled with one another.
 9. The filter circuit of claim 8 wherein eachof the first plurality of shunt acoustic resonators has a seriesresonance frequency that is different from the series resonancefrequency of the other of the first plurality of shunt acousticresonators.
 10. The filter circuit of claim 9 wherein each of the secondplurality of shunt acoustic resonators has a series resonance frequencythat corresponds to one of the first plurality of shunt acousticresonators.
 11. The filter circuit of claim 8 wherein a filter responseof one of a passband, a stopband, a multiple passband/stopband isprovided between the first node and the second node.
 12. The filtercircuit of claim 8 further comprising: a third node and a fourth node; athird plurality of acoustic resonators that are coupled in parallel withone another between the input node and the first node; a third inductorcoupled between the third node and the fixed voltage node such that thethird plurality of acoustic resonators is coupled in series with thethird inductor between the input node and the fixed voltage node; afourth plurality of acoustic resonators that are coupled in parallelwith one another between the fourth node and the output node; and afourth inductor coupled between the fourth node and the fixed voltagenode such that the fourth plurality of acoustic resonators is coupled inseries with the fourth inductor between the output node and the fixedvoltage node, wherein the third inductor and the fourth inductor arepositively coupled with one another.
 13. The filter circuit of claim 12wherein each of the first plurality of shunt acoustic resonators has aseries resonance frequency that is different from the series resonancefrequency of the other of the first plurality of shunt acousticresonators.
 14. The filter circuit of claim 13 wherein each of thesecond plurality of shunt acoustic resonators has a series resonancefrequency that corresponds to one of the first plurality of shuntacoustic resonators.
 15. The filter circuit of claim 14 wherein: each ofthe third plurality of shunt acoustic resonators has a series resonancefrequency that is different from the series resonance frequency of theother of the third plurality of shunt acoustic resonators; each of thefourth plurality of shunt acoustic resonators has a series resonancefrequency that corresponds to one of the fourth plurality of shuntacoustic resonators.
 16. The filter circuit of claim 15 wherein each ofthe first and second plurality of shunt acoustic resonators has a seriesresonance frequency that is different from the series resonancefrequency of each of the third and fourth plurality of shunt acousticresonators.
 17. The filter circuit of claim 16 wherein a filter responseof one of a passband, a stopband, a multiple passband/stopband isprovided between the first node and the second node.
 18. The filtercircuitry of claim 12 wherein a filter response of one of a passband, astopband, a multiple passband/stopband is provided between the firstnode and the second node.
 19. A differential power amplifier circuitcomprising: a first power amplifier having a first input node and afirst output node; a second power amplifier having a second input nodeand a second output node; a first transformer comprising: a primary sidecoupled between the first output node and the second output node; and asecondary side formed from a first inductor and a second inductorcoupled in series between a first node and a second node wherein acommon node is provided between the first inductor and the secondinductor; at least one series acoustic resonator coupled between thefirst node and the second node wherein at least one main seriesresonance is provided between the first node and the second node at amain resonance frequency through the at least one series acousticresonator; a first shunt acoustic resonator coupled between the commonnode and a fixed voltage node; and a second shunt acoustic resonatorcoupled between the common node and the fixed voltage node, wherein adifferential input is provided across the first input node and thesecond input node, and a single-ended output signal is provided at thesecond node.
 20. The differential amplifier circuit of claim 19 whereina first series resonance at a first resonance frequency and a secondseries resonance at a second resonance frequency, which is differentfrom the first resonance frequency and the main resonance frequency, areprovided between the first node and the second node.
 21. Thedifferential amplifier circuit of claim 19 wherein the first inductorand the second inductor are negatively coupled with one another.
 22. Thedifferential amplifier circuit of claim 19 wherein at least one of thefirst resonance frequency and the second resonance frequency is greaterthan the main resonance frequency.
 23. The differential amplifiercircuit of claim 19 wherein the first resonance frequency is less thanthe main resonance frequency, and the second resonance frequency isgreater than the main resonance frequency.
 24. The differentialamplifier circuit of claim 19 wherein the at least one series acousticresonator comprises a plurality of acoustic resonators that are coupledin parallel with one another and each of the plurality of acousticresonators has a different series resonance frequency.
 25. Thedifferential amplifier circuit of claim 19 further comprising at leastone additional shunt acoustic resonator coupled between the common nodeand the fixed voltage node.